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BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates
generally to headphones, and more particularly to a novel technique for reducing distortion and
providing noise while providing a relatively constant frequency response that is not altered by
the user. Devices and techniques. The invention achieves this feature by means of a relatively
compact headphone which can be worn comfortably without exerting excessive pressure on the
head by the force of pressing the cup on the head.
BACKGROUND OF THE INVENTION A typical prior art method of attenuating noise uses a
headphone with a high mass, large volume, spring-loaded device that exerts a large pressure on
the head. The high mass increases the inertia that resists acceleration and contributes to the
structural rigidity of the headphone wall. The high pressure has the effect of increasing the
damping of the closed low frequency without air leakage. Large volume compliant air cavities
provide high frequency roll off. However, most of these techniques increase discomfort for the
Conventional active noise cancellation techniques include methods of converting external noise
using a microphone external to the headphone. The electrical device then processes the
converted noise signal as well as the attenuation caused by the headphone to the noise signal to
provide the headphone driver with the reverse phased signal to cancel external noise Do. This
method may actually increase the noise level inside the headphone in an open loop system which
is not applicable to different users. Another approach is to use a closed loop or servo mechanism,
for example, "A STUDY OF PROPOSED EAR, by Patrick Michael Dalosta, April 1975, described in
report AB-A 009 274 distributed by the National Technical Information Service. PROTECTION
DEVICES FOR LOW FREQUENCY NOISE ATTENUATION. U.S. Pat. No. 3,009,991 discloses a speed
sensing microphone provided in close proximity to the diaphragm of the speaker in the feedback
loop. U.S. Pat. No. 3,562,429 discloses movable feedback, remote acoustic feedback and feedback
around the headphone.
SUMMARY OF THE INVENTION It is therefore an object of the present invention to provide a
headphone apparatus which ameliorates the aforementioned problems.
SUMMARY OF THE INVENTION In accordance with the present invention, there is provided
means for defining a headphone cavity and electroacoustic transducing means such as a pressure
sensitive microphone provided in the cavity, with external noise and in the same cavity. Provide a
signal corresponding to the sum with the sound generated by the headphone driver.
It also has means for combining the converted signal with the desired input signal to be
reproduced, the error signal representing noise and the difference between the input signal to be
reproduced and the output of the headphone driver in its cavity Generate. Servo means, including
combinatorial means, compensates for these error signals to provide the output acoustic signal to
the ear with significantly reduced external noise and distortion, and also the input and ear to
which the desired signal to be reproduced is applied. Give a nearly uniform frequency response
Cushioning means are provided between the ear and the headphone to prevent significant
leakage and to prevent appreciable divergence in the pressure field. The cushioning means is
sufficiently flexible, adapts to the shape of the irregular ear to provide the desired seal, has a
sufficiently high density and flow resistance, and the cavity portion defined has rigid walls Acting
as a nearly ideal cavity, the pressure wave at the cavity wall meets the zero velocity boundary
condition so that the pressure amplitude is approximately constant for wavelengths much larger
than the maximum distance across the cavity. The electroacoustic transducing means is arranged
to be responsive to the pressure in the cavity near the ear in proximity to the headphone driver
means. Preferably, the headphone driver means comprises a diaphragm operating as a piston in
the relevant audio frequency range. The span across the diaphragm is desirably small, eg, at its
diameter for circular diaphragms, eg, less than one-third wavelength at the highest operating
audio frequency.
EXAMPLES The present invention will be described in detail by way of examples.
Referring to FIG. 1, a schematic view of the headphone of the present invention attached to the
ear is shown.
The microphone 11 is disposed coaxially with the headphone housing 13, the driver 17 and the
driver diaphragm 14 in the cavity 12, and the cushion 15 seals the area between the outer ear
16 and the cushion support 21 of the headphone. The microphone 11 is disposed close to the
entrance of the ear canal 18 so that the amplitude of the pressure wave at the microphone 11 is
substantially the same as the entrance of the ear canal 18. The cavity 12 is reduced such that the
pressure is substantially constant throughout the cavity. To achieve this end, the cushion 15 has
high mechanical compliance, high flow resistance, high density, an axial cross section
approximately equal to the diaphragm 14 and an axial annular shape of the cushion 15 around
the cavity 12 Smaller than the cross sectional area. The headphone housing 13 is coupled to the
resilient headband 23 in a known manner by means of a frictional pole joint 22 as shown in FIG.
A typical material for the headphone cushion 15 is an open cell urethane foam (foam) that is
slow to recover. The cushion 15 holds the outer ear 16 over a relatively large area, distributes
and effectively seals the force necessary to maintain a good seal over a sufficiently large area,
and reduces the pressure on the ear sufficiently. To make the user uncomfortable. The open-cell
high-flow resistance material exhibits the mechanical advantage of the open-cell material that
conforms to the irregular shape of the ear while at the same time providing a closed-cell in that it
significantly attenuates the spectrum component of middle range predetermined frequency, for
example 2 KHz or more. It has the acoustic advantage of a (closed cell) material. Also, the
pressure in the cavity is maintained substantially constant in this frequency range. Fluid filled
cushions also have these properties. Such a configuration of the present invention is generated
overhead in response to the force required to hold a closed or open cell low flow resistance
cushion around the ear that attenuates the low frequency signal as in the prior art. This is
different from providing a large cavity with high pressure. Such conventional cavities are
characterized by divergence in a large pressure field, and a diaphragm deflection movement of a
larger diaphragm is required to generate a constant sound pressure level than the small cavities
according to the invention. In this way, better acoustic efficiency can be achieved with a small set
that gives the user of the present invention more comfort.
Referring to FIG. 2, a block diagram illustrating the logical configuration of the device according
to the present invention is shown. Signal combiner 30 algebraically combines the signal desired
to be reproduced by the headphone at input terminal 24 with the feedback signal provided by
microphone preamplifier 35. Signal combiner 30 supplies the combined signal to compressor 31,
which limits the level of the high level signal. The compressor then sends the compressed signal
to the compensator 31A. The compensation circuit 31A adapts the open loop gain to the Nyquist
stability criterion so that it does not oscillate when the loop is closed. The illustrated device is
made similarly for the left and right ears.
The power amplifier 32 energizes the headphone driver 17 to generate an acoustic signal in the
cavity 12, which signal is combined with an external noise signal entering the cavity from the
area of the acoustic input terminal 25 in a portion indicated by a circle 36, The combined sound
pressure signal is applied to the microphone 11 and converted there. The microphone
preamplifier 35 amplifies the converted signal and transmits it to the signal combiner 30.
The compensation circuit 31A is designed so that the loop gain T (s) is maximum (open loop) at
40 to 2000 Hz as shown by the curve 20 in FIG. This loop gain is represented by P (s) =
(CBDEMA), where DM is the transfer function of the electrical signal output of the microphone
11 with respect to the electrical input to the driver 17, and A, B, C, E, D and M is the transfer
characteristic of the microphone preamplifier 35, the power amplifier 32, the compressor circuit
31, the compensation circuit 31A, the driver 17 and the microphone 11, respectively. The loop
gain is maximized to allow for sufficient phase and amplitude margins to ensure stability under
different conditions, including head-to-head differences and head-offs.
The closed loop transfer function PO from the electrical input to the pressure output is PO / VI =
Tu = CBDE / (1 + CBDEMA) at the entrance to the ear canal. The amplitude of this closed loop
transfer function as a function of frequency corresponds to the curve 21 of FIG. The amount of
active noise reduction when PI corresponds to acoustic noise input is (PO / PI) = NR = 1 +
CBDEMA = 1 + T (s)
Referring to FIG. 3, a curve 23 representing the actual noise reduction measured by the middle
ear simulated microphone is shown relative to the theoretical curve 24 obtained by adding one to
the open loop gain. .
Referring to FIGS. 5 and 6, a block diagram illustrating the theoretical configuration of the
preferred embodiment of the present invention is shown.
For convenience, it is represented by six blocks indicated by numerals 1 to 6, the compensation
circuit block 2 is further divided into sub blocks 2a, 2b and 2c, and the compressor block 5 is
further divided into five subblocks 5a, 5b, 5c and 5d. And 5e. The circuitry forming the blocks of
FIGS. 5 and 6 is shown in FIGS. 7 and 8 at the boundaries of the dashed lines. A person skilled in
the art can implement the present invention by assembling the circuits of FIGS. 7 and 8, so a
detailed description will not be necessary. Next, an embodiment of the present invention will be
described with reference to FIGS. 5, 6, 7 and 8. FIG.
The adder / multiplier 1 has an adder 30 for receiving the input audio signal from the input
terminal 24 and the feedback signal supplied from the amplifier 35. The adder 30 is realized by
an operational amplifier connected in the form of a normal inverting summing amplifier as
shown in FIGS. 7 and 8. A capacitor connected between the connection point of a pair of similar
input resistors and the system ground shunts high frequency signal components to the system
ground. One half of the analog switch U304 transmits the remaining low frequency components
to the virtual ground at the input of the integrated circuit U102. The modulation signal on the
MOD line from the compressor block 5 turns the switch on and off at a rate of 50 KHz. The
switch closing time in each cycle is controlled to perform multiplication. For signals characterized
by spectral components of bandwidth much less than 50 KHz, the switch can be considered as an
impedance of magnitude proportional to the duty cycle of the modulated signal waveform. In
normal operation, the switch is closed for most of the time. When the compressor 5 detects a
very large input amplitude, the duty cycle is reduced to attenuate the low frequency spectral
content of the input signal.
The series connected resistor and capacitor transmit the high frequency signal component
directly to the input of U 102 so as not to be affected by the multiplication operation.
The output of the adder / multiplier block 1 is sent to a compensation block 2, which is
characterized by amplitude and phase characteristics that guarantee the stability of the feedback
loop without reducing the overall loop gain so much. • Composed of filters.
Section 2c provides gain with appropriate roll-off at high frequencies. Sections 2a and 2b
compensate for the phase response of the loop gain at low and high frequency crossover points,
respectively. The principle of this preferred form of compensation will be described later, but it
maintains stability to prevent oscillation while providing high gain in the frequency band that
contains most of the voice spectral components.
Power amplifier block 3 receives the output signal from compensation block 2. Section 3b is a
known non-inverting amplifier with a separate output current buffer. Section 3a is a simple diode
limiter that protects destructive power consumption levels from being applied to the driver. The
light emitting diode emits light when the limiter operation is occurring. Preferably the input is AC
coupled to remove the DC offset from the previous stage.
The driver / microphone / ear system block 6 is not shown in FIGS. 7 and 8. The driver 17
receives the amplified signal from the power amplifier block 3 and generates an acoustic signal
which is sensed by the ear 16 and converted by the microphone 11. Incomplete sealing of the
cushion 15 results in low frequency roll-off. Composite structures that produce multiple
resonances at frequencies above a few KHz are also features of block 6. Furthermore, the
propagation delay from driver 17 to microphone 11 and the distributed source characteristics of
driver 17 cause excessive phase shifts. However, the components of the system work together to
compensate for these non-uniform characteristics and produce a substantially uniform closed
loop frequency response between the input terminal 24 and the ear canal 18.
The microphone preamplifier block 4 is a low noise operational amplifier connected to be a noninverted gain that receives the converted signal from the microphone 11. The amplifier and gain
are chosen so that the microphone's self noise dominates, thereby minimizing the effect the
system electronics have on the noise level at the ear 16. The zener diode supplies a bias voltage
VCC to the electret microphone 11. The amplifier 35 of the adder / multiplier block 1 receives
the amplified signal provided by the microphone preamplifier 4.
The compressor block 5 monitors both the signal at the input terminal 24 and the feedback
signal at the output of the microphone preamplifier 4 and supplies a modulation signal to the
MOD line which modulates the low frequency gain of the adder / multiplier block 1 . Section 5a
adds both the left and right channel feedback signals and the input signal. A low pass filter with a
corner frequency (typically 400 Hz) selectively transmits the combined signal for full wave
rectification. Section 5b averages the rectified signal in fast attack and slow decay times and
provides an output signal proportional to the low frequency spectral energy of both the left and
right loops. Section 5c converts the latter signal into a proportional current with offset, whose
gain and offset are controlled by a potentiometer.
Section 5d receives the output current signal from section 5c and provides a 50 KHz modulation
signal on the MOD line. Integrated circuit U 305 of FIG. 7 consists of a 50 KHz clock pulse source
that triggers integrated circuit U 306 and resets its output to ground every 20 microseconds.
Linearly falling to a threshold level at a rate proportional to the output current provided by the
capacitor voltage section 5c of pin 2 of integrated circuit U306, switching the output switch of
integrated circuit U306 high, resetting the capacitor voltage at terminal 2 And the positive
supply voltage to be triggered again. Since the analog switches of adder / multiplier block 1 are
closed to the ground potential of control pins 1 and 8, the adder / multiplier gain for low
frequencies is inversely proportional to the current level provided by section 5c. The large
current causes the capacitor potential of pin 2 of integrated circuit U306 to reach the threshold
level at high speed, and analog switch U304 is correspondingly closed in a short time. Section 5e
drives an LED bar graph display that displays the amount of compression. Since low frequency
spectral components transmit most of the typical input signal, it is sufficient to sense low
frequencies and operate accordingly accordingly.
In summary, the system can be thought of as a servo system having two input signals. The first
input is the audio electrical signal to be reproduced. The second input is an acoustic noise signal
around the ear. The output of the system is an acoustic signal generated in the ear. The feedback
signal is a voltage proportional to the instantaneous sound pressure at the entrance to the ear
canal. This sound pressure is a combination of the sound supplied by the driver and the
surrounding acoustic noise. The small electret microphone 11 converts this signal, the preamp
block 1 amplifies it, and the adder / multiplier block 1 adds this feedback signal to the input
audio signal to get the actual sound pressure to the ear An error signal is provided that
represents the difference from the desired sound pressure. The desired sound pressure is
proportional to the input audio signal. The compensation block 2 selectively transmits the
spectral components of the error signal to ensure loop stability. The amplifier block 3 amplifies
the compensated signal and sends it to the driver 17 to generate at the ear a sound pressure
corresponding to the desired audio input signal. Thus, over the frequency range in which the
feedback loop is active, the loop corrects for the spectral coloration of the driver / microphone /
ear system to cancel ambient noise. The amount of correction is related to the amount of stable
gain that the loop can supply. The compressor block 5 cooperates with the multiplier portion of
the adder / multiplier block 1 to prevent the input audio or audio signal from overdriving and
clipping the loop.
Having described the preferred embodiment of the present system, subsystems and their features
will now be described. The compressor block 5 is implemented in a particularly advantageous
manner which reduces artifacts during compression and avoids non-linear oscillations.
Conventional compressors are typically classified into basic types, n to 1 compression and
threshold compression. The n to 1 compressor produces a 1 dB change in output level for each
ndb input level. The threshold compression is linear with respect to an input signal equal to or
less than a threshold, and when this threshold is exceeded, an n to 1 compression ratio is
obtained, and this ratio may become infinite above the threshold level, which limits the average
output level.
The n to 1 compressor compresses the signal transmitted through the noisy communication
channel or the signal recorded on the noisy medium, and then decompresses the compressed
signal after detection so that the recording or communication channel noise is decompressed.
Widely used in companders that restore to their original dynamic range with significant damping.
Threshold compressors are commonly used in systems where the signal is not later
decompressed because, when properly designed, there are fewer unwanted artifacts in the output
signal than an n to 1 compressor.
A threshold compressor with an infinite compression ratio above the threshold is typically
created with a feedback loop. If the compressor gain and compressor attack and decay time
constants are not carefully selected, significantly undesirable audible sounds may be generated,
especially for input levels just above the threshold, causing the system to oscillate, Can produce
unpleasant audible sounds.
The present invention is an improvement over conventional threshold compressors that have
infinite compression ratios above the threshold. FIG. 9 is a block diagram showing the logical
configuration of the device implemented in block 5. This device responds to the input signal X at
terminal 51 and supplies a compressed signal Y at terminal 52. The dividend input of the divider
53 is connected to the input terminal 51. The input terminal 51 is also connected to the input of
the full wave rectifier 54. The output of the full wave rectifier 54 is connected to the input of the
averaging low pass filter 55, which is characterized by a decay response time constant which is
much larger than the attack response time constant. The output (X) of the averaging low pass
filter 55 is connected to the input of an amplifier 56 having a compressor gain K. Adder 57
receives signal K0 at one of its inputs and provides a divisor signal at the divisor input of divider
53. The divider 53 provides the quotient signal X / a at the input of the output amplifier 58
which supplies the compressed output signal Y.
It can be seen that for static signals, for example sinusoidal waves, the input-output gain is Y / X
= 1 / (K0 + K (X)). FIG. 10 is a graph showing this compression characteristic. This characteristic
is an infinite compression ratio above the threshold where Y / X = 1 / K0 = constant for small
signals and Y / X = 1 / KX or Y = 1 / K for large signals Similar to the characteristics of the
threshold compressor with. However, the compressor according to the invention has at least two
advantages as compared to conventional compressors. The transition from uncompressed to full
compression is smooth, so there is less audible compression generated for complex input signals,
such as music. Furthermore, since there is no feedback, nonlinear oscillation can not occur.
Here, a preferred compression mode will be described. If the attenuation of the gain A (ω) is
known for the entire frequency range, then the phase φ (ω) for the minimum phase network is
unambiguously determined, and likewise φ (ω) is known for the entire frequency range. When
being A (.omega.) Is unambiguously determined for a "minimum phase" function without poles
and zeros in the right half of the s or p plane. This property, which is already known, is M.I. I. T.
RADIATION LABORATORY SERIES vol. 25 Theory of Servomechanisms , 4.9 4.9
relationship was first described by Y. in Journal of Mathematics and Physics, June 1932. W. As
Van Nostrand Co. New York, 1945), chapter XIV, "Relations between Real and Imagimary
Composition of Network Functins". In 4.8 4.8 of the "Theory of Servomechanisms" mentioned
above, the "damping-phase" type analysis is described as the most satisfactory approach to servo
design problems, and the phase margin criteria at the feedback cutoff frequency are A good
practical measure of system stability is stated to be at least 30 °, preferably 45 ° or more. For
cut-off frequencies above 6 db per octave this phase should have sufficient phase margin for
frequencies with a gain A of 1 (log A = 0) at least 2 1/2 octaves above the cut-off frequency. It is
described that there is.
The purpose of providing a phase margin is to avoid situations that may cause unwanted
oscillations to continue and to eliminate peaking that amplifies external noise. The disadvantage
of providing such a large area between the frequency fc of unity and the corner frequency is that
the desired effect of negative feedback on spectral components within that frequency range is
significantly reduced. The present invention gives this phase the appropriate phase margin at
unity gain frequency fc by combining the networks with high open loop gain while maintaining
stability in the relevant frequency band. To establish both attenuation or gain characteristics and
phase characteristics.
The present invention establishes a stable phase margin at frequency fc where the gain drops to
zero while at the same time compensation characterized by an open loop gain or damped
frequency response with arbitrary slope at the passband or multiple ends. Includes means. These
principles are further understood by the examples described below.
Referring to FIGS. 11 (a) and 11 (b), graphs of gain or attenuation and phase characteristics of an
operational amplifier having normal first order characteristics are shown. The gain is constant up
to the half power or break frequency ω 0 and then decreases linearly at 6 db per octave.
Referring to FIGS. 12 (a) and 12 (b), the attenuation and phase characteristics shown in FIGS. 11
(a) and 11 (b) are modified by applying the principles of the present invention to obtain
substantially the same phase margin. A graph is shown that achieves greater loop gain above f0
while maintaining. This is accomplished by adding a dashed line to the break point at f1 above f0
and providing a slope of 12 db per octave. The corrected phase characteristic shown by the
broken line in FIG. 12B has a phase margin of approximately π / 2.
Referring to FIG. 13, a modified bandpass response is shown with dashed lines representing the
gain or attenuation correction for a more conventional approach having a slope of 6 db per
octave on either side of the transmission band. These compensation circuits have a slope close to
the break frequency of the amplitude that is greater than the slope at a frequency close to the
critical frequency fc of unity, while having a suitable phase margin.
The compensation circuit of block 2 of FIGS. 5, 6 and 7 and 8 implements these principles. It can
be seen that the loop compensation follows the guidelines from the open loop gain curve. In FIG.
4, the slope after the corner frequency (500 Hz) is 18 db / octave. The network of section 2a has
a zero at 80 Hz, a pole at 92 Hz, and damping constants of 0.56 and 1.1, respectively. The high
frequency circuit of section 2b has a zero at 3.1 KHz and a pole at 7.3 KHz, and has damping
constants of 0.49 and 1, respectively. The circuit of section 2c and the low pass filter have zeros
at 1.6 KHz and 3.4 KHz and poles at 160 Hz, 320 Hz, 800 Hz and 34 KHz.
Although the present invention has been described above according to the embodiments, it is
obvious to those skilled in the art that many variations and modifications are possible within the
scope of the present invention.
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